1. Field of the Invention
The invention relates to a wireless communication system, and more particularly, to a receiver of the wireless communication system.
2. Description of the Prior Art
Orthogonal Frequency Division Multiplexing (OFDM) is a multi-carrier transmission scheme used in high-speed data communications (e.g., both the IEEE 802.11a and 802.11g wireless LAN standards use OFDM). An OFDM transmitter utilizes a plurality of tones, also known as sub-carriers, which are sinusoidal waves and are orthogonal to one another. Each tone carries a certain bit-load of information using a certain modulation scheme, such as binary phase shift key (BPSK) that carries 1-bit of information, quadrature phase shift key (QPSK) that is also known as 4-QAM and carries 2-bits of information, 16-points quadrature amplitude modulation (16-QAM) that carries 4-bits of information, 64-points quadrature amplitude modulation (64-QAM) that carries 6-bits of information, and so on.
FIG. 1 shows a block diagram of a typical OFDM transmitter 100. In FIG. 1, N tones are utilized for transmission (labeled 0 to N−1). The input serial data is buffered and converted into N parallel data, labeled 0 to N−1 according to the respective bit at the serial in, parallel out (SIPO) buffer 101. A BPSK/QPSK/QAM mapper 102 encodes each of the N parallel data paths into a respective complex number, which represents the in-phase and quadrature components for its respective tone. An N-points Inverse Fast Fourier Transform (IFFT) module 104 is used to generate N time-domain complex samples (labeled 0 to N−1). The last NGI samples of the IFFT output are pre-pended to the beginning of the IFFT output. This process is referred to as “circular prefix” and is used to form a guard interval. Note that the guard interval is used to provide a time-domain buffer between adjacent OFDM symbols so as to improve the receiver's immunity to ISI (inter-symbol interference). The resultant (N+NGI) samples form an OFDM symbol, consisting of the superposition of N tones in time domain, each tone carrying its respective information. These samples are converted into serial time-domain samples in the subsequent parallel in, serial out (PISO) buffer 108.
The (N+NGI) serial time-domain samples are complex data, which can be separated into an in-phase component (real part) and a quadrature-phase component (imaginary part). Both components are converted to an analog signal at their respective digital-analog converter (DAC) 112, 118, resulting in an in-phase signal I(t) and a quadrature signal Q(t). A local oscillator (not shown) generates a quadrature carrier pair, cos(wt) and −sin(wt), which are used to modulate I(t) and Q(t), respectively. The resultant RF signals are summed, amplified, and transmitted to the receiver via the channel, which could be either a wireless medium or a transmission line.
FIG. 2 shows a block diagram of a typical OFDM receiver 200. As shown in FIG. 2, the received signal is amplified by a low noise amplifier 202, separated into I-path and Q-path, and then demodulated by a quadrature carrier pair (i.e., cos(wt) and −sin(wt)). Low pass filtering (LPF) is performed on both the in-phase and quadrature-phase paths to provide an anti-aliasing function before the I-Q signals are converted into digital samples by a pair of analog to digital converters (ADCs) 208, 214. The serial time domain samples obtained from the ADCs 208, 214 form a sequence of serial complex samples. The serial complex samples are buffered and converted by a SIPO 220 into (N+NGI) parallel samples, labeled 0 to (N+NGI−1) in FIG. 2. The guard interval samples are removed, resulting in N complex samples, again labeled 0 to (N−1). A subsequent N-points Fast Fourier Transform (FFT) 224 generates N frequency domain samples from the N time-domain samples. The frequency domain samples are adjusted in both magnitude and phase by a subsequent frequency-domain equalizer (FEQ) 226 to compensate for the amplitude change and phase shift caused by the channel on a per-tone basis. A per-tone decision on each of the N FEQ outputs is made in the subsequent BPSK/QPSK/QAM demapper 228, resulting in N parallel data paths, where each data path carries a certain bit-load of information. The N parallel data are then converted into serial data by a subsequent PISO 230.
In practice, the OFDM receiver 200 suffers from a phenomenon known as the quadrature mismatch (or I-Q mismatch) problem. As shown in FIG. 3, the quadrature carrier pair generated by a local oscillator 300 within the receiver exhibits a slight phase mismatch. Therefore, instead of generating cos(wt) and −sin(wt) differing perfectly by 90 degrees, the local oscillator 300 generates cos(wt) and −sin(wt+δ). The circuit blocks on the I-path (LPF1 206 and ADC1 208) will also be slightly different from those on the Q-path (LPF2 212 and ADC2 214), as they are all subject to limited component tolerances. These differences and the non 90 degree phase difference both result in I-Q mismatch. To account for the I-Q mismatch, some compensation needs to be performed within the receiver 200.
FIG. 4 shows a typical I-Q mismatch correction circuit 400 according to a first correction scheme of the related art. An amplitude mismatch estimate block 404 is used to estimate the amplitude mismatch between the I and Q paths and to accordingly adjust the gain of one of the paths (e.g., the Q-path in this example) to thereby compensate for the amplitude mismatch. A phase mismatch estimate block 406 is used to estimate the phase mismatch between the I and Q paths and to accordingly adjust the delay on one of the quadrature carrier pair (e.g., again the Q-path in this example) to thereby compensate for the phase mismatch.
FIG. 5 shows another I-Q mismatch correction circuit 500 according to a second correction scheme of the related art. In this circuit 500, instead of physically adjusting the phase of the quadrature carrier pair, both the amplitude mismatch A and the phase mismatch a are estimated. One path (e.g., the Q-path in this example) is scaled by a factor of G2=1/[A×cos(a)] to correct the slight amplitude reduction due to amplitude and phase mismatch, and the cross coupling contribution from this path (i.e., the coupling factor of G2=tan(a) is subtracted from the other path (i.e., the I-path in this example). In doing so, the distortion due to phase mismatch, which is basically I-Q coupling, is removed.
However, the prior art I-Q mismatch correction schemes of FIG. 4 and FIG. 5 have the drawback that they cannot effectively correct I-Q mismatch if the mismatch is frequency dependent. Frequency dependency of the I-Q mismatch problem is particularly pronounced in wide-band applications where the circuit elements need to process a wide range of frequency components. Therefore, an improved method of correcting I-Q mismatch in an OFDM receiver would be desirable.